CN112366936A - Low-output ripple power factor correction converter - Google Patents

Abstract

The invention discloses a low-output ripple power factor correction converter, which specifically includes a rectifier bridge, a filter, a Buck PFC converter, a DC-DC converter in a subsequent stage, and a control circuit thereof; the filter consists of L _f and C _f ; the transformer is equivalent to the form of excitation inductance, ideal transformer and leakage inductance; the Buck PFC converter and the subsequent DC-DC converter are integrated by sharing the active switch; the Buck PFC converter is composed of diodes, inductors , capacitors and active switches; DC‑DC converters consist of diodes, resonant capacitors, output capacitors, transformers, active switches and energy storage capacitors. The invention has the advantages of high efficiency, high power factor, low output ripple, and easy expansion into multiple outputs. In addition, the present invention is an isolation structure, which is more secure.

Description

A Low Output Ripple Power Factor Correction Converter

technical field

The invention belongs to the technical field of converters, and in particular relates to a low output ripple power factor correction converter.

Background technique

Power factor correction (PFC) converters are widely used to reduce input current distortion and meet relevant harmonic standards. In order to reduce the impact of power electronic equipment on the power quality of the power grid, the international IEC61000-3-2 Class C and the national harmonic standard GB/T 14549-1993 "Power Quality Public Power Grid Harmonics" have an impact on the PF (Power factor) of power electronic equipment. , power factor) have strict requirements. Therefore, it is of great significance to use an LED driver with PFC function.

Generally, PFC converters can be classified into single-stage, two-stage and quasi-single-stage structures. The main advantages of the single-stage structure are high efficiency and small size. However, no matter what topology or control method is used, there is always an instantaneous energy imbalance between the AC input and the DC output of the single-stage structure, resulting in a large output. double the power frequency ripple. Especially in the application of LED driver, large ripple current will cause LED stroboscopic, which will lead to a series of health problems such as headache and eye fatigue.

The two-stage structure is usually composed of the front-stage PFC converter and the latter-stage DC-DC converter for eliminating the double power frequency ripple. The two-stage structure can achieve high power factor and meet relevant harmonic requirements. However, the two-stage structure requires at least two active switches and two control loops, which reduces the efficiency of the converter and increases the cost and control complexity of the converter. In recent years, many different quasi-single-stage topologies have been introduced in many literatures to improve converter efficiency and reduce output current ripple. However, compared with the single-stage structure, the quasi-single-stage structure still requires more than two active switches, and the control is more complicated.

In addition to the above topologies, many studies have been devoted to combining the advantages of single-stage and two-stage structures, usually by integrating the PFC converter unit and the DC-DC converter unit through an active switch to form an integrated topology. The integrated topology reduces the number of components and costs compared to the standard two-stage topology. At the same time, the integrated structure can also achieve low output ripple and high power factor. Moreover, the integrated structure is much simpler to control than the standard two-stage structure and quasi-single-stage structure. However, the main disadvantage of the integrated structure is that the voltage stress of the common power switch is too high, which limits its efficiency improvement.

In the prior art, as shown in FIG. 1, a high power factor multi-channel low ripple constant current output switching converter is an integrated Boost-Buck converter. While achieving low output current ripple, only one One set of control of switch tubes has lower cost and higher efficiency; while realizing power factor correction, the multi-channel current sharing output of LED driving power is realized by using capacitor charge and discharge balance, which reduces the volume and size of the lighting system. cost, which is a great advantage in applications that require multiple outputs. It is characterized in that, as an n-channel LED driving power supply, it includes a diode rectifier bridge D _b , an input filter inductor L _f , an input filter capacitor C _f , a main inductor L , an active switch S ₁ , a sampling resistor R _s , and each branch. The storage capacitors C ₁ , C ₂ , ..., C _{(2 n -3)} , C _{(2 n -2)} in the output light-emitting diode load LEDs ₁ , LEDs ₂ , ..., LEDs _n , the freewheeling diode D ₁ , D ₂ , D ₃ , ..., D _{(2 n -2)} , D _{(2 n -1)} , output capacitors C _o1 , C _o2 , ..., C _{0 n} and branch inductances L ₁ , L ₂ , ..., L _n . The current and voltage parameters in the figure are: i _in is the input current, i _o1 , i _o2 ,..., i _on are the current of the first branch, the current of the second branch,..., the nth branch respectively current. The control loop includes _: a proportional integral link composed of a resistor Re , a capacitor C _e and an error amplifier EA, a comparator COMP and a drive. In the control loop, v _saw is the _sawtooth wave signal, V _ref is the reference voltage, and ve is the error signal output by the error amplifier.

The control loop adopts a wide bandwidth voltage loop to control the inductor current of the energy storage inductor L to work in discontinuous mode, and the inductor current of the branch inductor works in discontinuous mode or critical continuous mode; energy storage capacitors C ₁ , C ₂ , ... , C _{(2 n -3)} , C _{(2 n -2)} The influence of the power frequency ripple on the output can be suppressed by a wide bandwidth voltage control loop.

Figure 2 shows the circuit topology and control circuit of the two-way Boost-Buck LED drive power supply. It is characterized in that, as a 2-channel LED driving power supply, it includes a diode rectifier bridge D _b , an input filter inductance L _f , an input filter capacitor C _f , a main inductance L , a sampling resistor R _S active switch S ₁ , and an active switch S 1 in each branch. energy storage capacitors C ₁ , C ₂ , freewheeling diodes D ₁ , D ₂ , D ₃ , output light-emitting diode loads LEDs ₁ , LEDs ₂ , output capacitors C _o1 , C _o2 and branch inductances L ₁ , L ₂ . The control loop includes _: a proportional integral link composed of a resistor Re , a capacitor C _e and an error amplifier EA, a comparator COMP and a drive. The current and voltage parameters in the figure are: i _in is the input current, v _in is the input voltage, | i _in | is the current after full-wave rectification of the input current, i _Lm is the current flowing through the inductor L , and v _s1 is S ₁ The voltages at both ends, i _L1 , i _L2 are the currents of the inductors L ₁ and L ₂ respectively, i _D2 is the current flowing through D ₂ , i _C1 , i _C2 are the currents flowing into the capacitors C ₁ and C ₂ respectively, v _C1 and v _C2 are the voltages of the output capacitors C ₁ and C ₂ respectively, v _o1 and v _o2 are the voltages of the output capacitors C _o1 and C _o2 respectively, i _o1 and i _o2 are the current of the first branch and the second branch respectively current. In the control loop, v _saw is the _sawtooth wave signal, V _ref is the reference voltage, and ve is the error signal output by the error amplifier. Figure 3 is the PSIM simulation waveform diagram of the input voltage and input current of the driving power supply, in the figure, v _in is the input voltage waveform, i _in is the input current waveform, the abscissa is the simulation time, the unit is second (s), and the ordinate is the input voltage The value scaled down by a factor of 100, in volts (V), and the value of the input current, in amps (A). Figure 4 is the simulation waveform diagram of the output current when the output load resistance is 150Ω and 150Ω respectively, i _o1 is the current of one output branch, i _o2 is the current of the other output branch, the abscissa is the simulation time, the unit is Second (s), the ordinate is the magnitude of the current, in ampere (A). Fig. 5 is the enlarged waveform of Fig. 4. Since the magnitudes and amplitudes of the two output currents i _o1 and i _o2 are almost equal, the waveforms of i _o1 and i _o2 in Fig. 5 overlap.

Its main features are:

1) The current sharing output of each branch is realized through the energy storage capacitor with special connection structure, and the current sharing accuracy is high; the circuit adopts a special output _current sampling method, and the power loop is different from the control loop; The output current of one branch is controlled to make its ripple smaller, and other output branches correspondingly realize low current ripple output; it has the characteristics of simple control and low cost;

2) The use of wide bandwidth voltage loop control reduces the influence of energy storage capacitor ripple on the output, realizes low ripple current output, and solves the problem of LED stroboscopic;

3) Based on the integrated Boost-Buck converter, while achieving low output current ripple, only one switch is required for a set of control, which has lower cost and higher efficiency;

4) While realizing the power factor correction, the multi-channel current sharing output of the LED driving power supply is realized by using the capacitor charge and discharge balance, which reduces the volume and cost of the lighting system, and has great advantages in applications that require multi-channel output. .

However, in this scheme, the front stage is a Boost converter, and the voltage stress of the switch tube is large. n outputs need n inductors, which increases the volume of the converter. In addition, the LED driver is non-isolated and cannot guarantee user safety.

SUMMARY OF THE INVENTION

To solve the above problems, the present invention provides a low output ripple power factor correction converter.

A low-output ripple power factor correction converter of the present invention includes a rectifier bridge, a filter, a BuckPFC converter, a post-stage DC-DC converter, and a control circuit thereof; specifically: an input filter inductance L _f and an input The filter capacitor C _f is connected in series and parallel to the output end of the rectifier bridge D _b ; one side of the input filter capacitor C _f is connected between the input filter inductor L _f and the Buck inductor L _B , and the other side is connected to the lower part of the diode rectifier bridge D _b . Output terminal; diodes D _B1 and D _B3 are connected in series and parallel to both ends of the input filter capacitor C _f , the anode of D _B3 is connected to the lower output terminal of D _b , and the cathode of D _B1 is connected between L _f and Buck inductance L _B ; the middle energy storage The positive stage of the capacitor C _B is connected to one end of the Buck inductor L _B , and the negative electrode is connected between D _B1 and D _B3 ; the anode of the diode D _B2 is connected to the negative electrode of the intermediate energy storage capacitor C _B , and the cathode is connected to the drain of the active switch S ₁ ; One end of the primary side of the transformer T is connected to the positive stage of the intermediate energy storage capacitor C _B , and the other end is connected to the drain _of the active switch S1 and the source of the active switch S2 _; the active switch S2 and the energy storage capacitor _{Cc are} connected in series It is then connected in parallel to both ends of the transformer T, wherein the drain of the active switch S ₂ is connected to the energy storage capacitor C _c , and the gate is connected to the control loop; the source of the active switch S ₁ is connected to the lower output end of the diode rectifier bridge D _b , the drain is connected to the source of S2 , and the gate is connected to the control loop _; one end of the secondary side of the transformer T is connected to the anode of D _o1 and the cathode of D _o2 , and the other end is connected to two series resonant capacitors C _r1 and C _r2 The cathode of D _o1 is connected to the resonant capacitor C _r1 and the output capacitor C _o ; the resonant capacitor C _r1 and C _r2 are connected in series and connected to the cathode of D _o1 and the anode of D _o2 ; the output capacitor C _o is connected in parallel with the resonant capacitor C The two ends after _r1 and C _r2 are connected in series; the output load is connected to both ends of the output capacitor C _o .

Further, by adding a diode and two output capacitors, the single-channel output can be expanded to three-channel current-sharing output, specifically: the primary circuit of the transformer T remains unchanged, and one end of the secondary side of the transformer T is connected to the D _o1 . The anode and the resonant capacitor C _r1 , the other end of the resonant capacitor C _r2 ; the other end of the resonant capacitor C _r1 is connected to the anode of D _o3 and the cathode of D _o2 ; the positive pole of the output capacitor C _o1 is connected to the cathode of D _o3 , and the negative pole is connected to the secondary of the transformer T The other end of the side, the output load R _o1 is connected to both ends of the output capacitor C _o1 ; the positive electrode of the output capacitor C _o2 is connected to the resonant capacitor _Cr2 , the negative electrode is connected to the anode of D _o2 , and the output load R _o2 is connected to both ends of the output capacitor C _o2 ; The negative pole of the output capacitor C _o3 is connected to the resonant capacitor C _r2 , the positive pole is connected to the cathode of D _o1 , and the output load R _o3 is connected to both ends of the output capacitor C _o3 .

The two active switches adopt _a complementary conduction mode, specifically: S2 is turned off before S1 is turned _on , and _both active switches S1 and S2 are turned off at this time _; S1 is turned on after _a short dead zone. , S2 remains off _; when S1 receives the shutdown signal, it is immediately turned off, and S2 continues to be off _; after _a short dead zone, S2 is turned _on , and S1 _remains off. The dead time is generated by the driver chip IR2111, specifically: the control signal generated by the control circuit is input to the driver chip IR2111, and then the IR2111 independently generates two complementary and symmetrical driving signals with dead zones to drive the active switches S ₁ and S respectively. ₂ .

The secondary side control realizes constant voltage/constant current control, specifically: one is the output voltage error amplifier EA1, and the other is the output current error amplifier EA2. It can be known from the working principle of the control loop that when the output voltage is lower than the controlled target voltage of the constant voltage output loop, the error output u _e1 of EA1 is high level, at this time D _c1 is reversely cut off, and the constant voltage output control loop is Failure, the constant current output error amplifier EA2 works normally, and the output is constant current control; on the contrary, when the output current is lower than the target control current of EA2, the error output of EA2 is high level, at this time, D _c2 is reversely cut off, and the constant current output The control loop fails, the constant voltage output error amplifier EA1 works normally, and the output is constant voltage control. Through the above control, constant current/constant voltage control can be realized.

The beneficial technical effects of the present invention are:

1. The traditional single-stage PFC converter has a large double power frequency ripple in its output voltage. If this kind of converter is applied to the LED occasion, the large double power frequency ripple will cause the strobe of the LED, which will lead to a series of health problems such as headache and eye fatigue in the human body. The output current ripple of the traditional two-stage PFC LED driver is very small, but because it requires two sets of control systems, the driver is large in size, high in cost and low in efficiency. The converter proposed by the present invention requires only one set of controls while realizing low output current ripple, and has lower cost and higher efficiency.

2. Traditional PFC converters are often only for a single load. For applications that require multiple current sharing outputs, multiple drivers are often required, and accurate current sharing cannot be guaranteed. Furthermore, traditional multi-output LED drivers are mostly based on half-bridge, full-bridge or DC-DC topology, and do not consider the PFC function, and often require additional PFC correction circuits, which increases the cost and volume of the system. While realizing PFC correction, the present invention is easy to expand into multi-channel current sharing output, reduces the volume and cost of the lighting system, and has great advantages in applications requiring multi-channel output.

3. The active switch shared by the traditional integrated structure has large voltage stress and voltage spikes. This reduces the efficiency of the converter and large voltage spikes tend to damage the active switches. The present invention utilizes the active clamp circuit to effectively reduce the voltage stress of the active switch and eliminate the voltage spike. At the same time, the active clamp circuit can recover the leakage inductance energy of the transformer and further improve the efficiency of the converter.

Description of drawings

Figure 1 is a schematic diagram of the Boost-Buck n-way LED drive power supply.

Figure 2 shows the circuit topology and control circuit of the two-way Boost-Buck LED drive power supply.

FIG. 3 is a PSIM simulation waveform diagram of the input voltage and input current of the driving power supply.

Figure 4 shows the simulation waveforms of the output current when the output load resistances are 150Ω and 150Ω respectively.

FIG. 5 is an enlarged waveform of FIG. 4 .

FIG. 6 is a low output ripple power factor correction converter of the present invention.

FIG. 7 shows the converter of the present invention expanded into multiple outputs.

Figure 8 shows the rectified input voltage and current waveforms.

Figure 9 shows the steady-state waveform of the converter under the condition of θ < ωt < π- θ (interval I).

Figure 10a shows the equivalent circuit of the converter operating in mode 1 under the condition of θ < ωt < π- θ (interval I).

Figure 10b shows the equivalent circuit of the converter operating in mode 2 under the condition θ < ωt < π- θ (interval I).

Figure 10c shows the equivalent circuit of the converter under the condition of θ < ωt < π- θ (interval I) when mode 3 is used.

Figure 10d shows the equivalent circuit of the converter operating in mode 4 under the condition θ < ωt < π- θ (interval I).

Figure 10e shows the equivalent circuit of the converter operating in mode 5 under the condition θ < ωt < π- θ (interval I).

Figure 10f shows the equivalent circuit of the converter operating in mode 6 under the condition θ < ωt < π- θ (interval I).

Fig. 10g shows the equivalent circuit of the converter working under the condition of θ < ωt < π- θ (interval I) when mode 7 and mode 9.

Figure 10h shows the equivalent circuit of the converter working in mode 8 under the condition θ < ωt < π- θ (interval I).

Figure 10i shows the equivalent circuit of the converter operating in mode 10 under the condition θ < ωt < π- θ (interval I).

Figure 10j shows the equivalent circuit of the converter operating in mode 11 under the condition θ < ωt < π- θ (interval I).

Figure 10k shows the equivalent circuit of the converter working in mode 12 under the condition θ < ωt < π- θ (interval I).

Figure 11 _shows the relationship between i _LB and id .

Figure 12 shows the steady-state waveforms of the converter under the conditions of 0 < ωt < θ and π-θ < ωt < π (interval II).

Figure 13a shows the equivalent circuit of the converter operating in mode 1 under the conditions 0 < ωt < θ and π- θ < ωt <π (interval II).

Figure 13b shows the equivalent circuit of the converter operating in Mode 2 under the conditions 0 < ωt < θ and π- θ < ωt <π (interval II).

Figure 13c shows the equivalent circuit of the converter operating in mode 3 under the conditions 0 < ωt < θ and π- θ < ωt <π (interval II).

Figure 13d shows the equivalent circuit of the converter operating in mode 4 under the conditions 0 < ωt < θ and π- θ < ωt <π (interval II).

Figure 14 shows the Buck inductor current and the rectified input voltage input current waveform.

Figure 15 shows power factor and efficiency.

Figure 16 shows the input current harmonic test results.

Detailed ways

The present invention will be further described in detail below in conjunction with the accompanying drawings and specific implementation methods.

A low-output ripple power factor correction converter of the present invention includes a rectifier bridge, a filter, a BuckPFC converter, a DC-DC converter at the rear stage, and a control circuit thereof. As shown in Fig. 6, the transformer _T is equivalent to the form of the excitation inductance Lm , the ideal transformer and the leakage inductance Lr _. 1: n is the number of turns on the primary side of the transformer compared to the number of turns on the secondary side. The Buck _- type PFC converter is integrated with the DC-DC converter of the subsequent stage by sharing the active switch S1 . Buck type PFC converter consists _of diodes ( DB1 , DB2 and _{DB3 )} , inductor _LB , capacitor _CB and active switch _S1 . The DC-DC converter consists of diodes ( D _o1 and D _o2 ), resonant capacitors ( C _r1 and C _r2 ), output capacitor C _o , transformer T, active switch S ₁ , energy storage capacitor C _c and active switch S ₂ composition. Specifically: the input filter inductor L _f and the input filter capacitor C _f are connected in series and then connected in parallel to the output end of the rectifier bridge D _b ; one side of the input filter capacitor C _f is connected between the input filter inductor L _f and the Buck inductor L _B , and the other side is connected between the input filter inductor L f and the Buck inductor LB The side is connected to the lower output terminal of the diode rectifier bridge D _b ; the diodes D _B1 and D _B3 are connected in series and then connected to both ends of the input filter capacitor C _f in parallel, the anode of D _B3 is connected to the lower output terminal of D _b , and the cathode of D _B1 is connected to L _f and Between Buck inductors LB; the positive stage of the intermediate energy storage capacitor C B _is connected to one end of the Buck inductor LB , and the negative electrode is connected between _{D B1} and D _B3 ; the anode of the diode D _B2 is _{connected to the negative electrode of the intermediate energy storage capacitor C B} , _and the cathode is connected Connect the drain of the active switch S1 ; _one end of the primary side of the transformer _{T is connected to the positive stage of the intermediate energy storage capacitor CB} , and the other end is connected to the drain _of the active switch S1 and the source of the active switch S2 _{; The} switch S2 and the energy storage capacitor Cc are connected in series and parallel to both ends of the transformer T. _The drain of the active switch S2 is connected to the energy storage capacitor _{Cc ,} and the gate is connected to the control loop; the source _of the active switch S1 is connected to To the lower output terminal of the diode rectifier bridge D _b , the drain is connected to the source of S2 , and the gate is connected to the control loop _; one end of the secondary side of the transformer T is connected to the anode of D _o1 and the cathode of D _o2 , and the other end is connected to the two terminals. Between two resonant capacitors C _r1 and C r2 in series; the cathode of Do1 is connected to the resonant capacitor C r1 and the output capacitor C o ; the resonant capacitors C r1 and C _{r2 are} connected _in series _and connected _to the _{cathode of} D _o1 and the anode of D o2 ; The output capacitor C _o is connected in parallel with the two ends of the resonant capacitor C _r1 and C _r2 after being connected in series; the output load is connected to both ends of the output capacitor C _o .

The current and voltage parameters in Figure 6 are: v _in is the input voltage, i _in is the input current, | i _in | is the current after full - wave rectification, i _LB is the current in the inductor LB, and i _B is the _inflow capacitor C _B i _d is the current flowing into the DC-DC unit, _ic is the current flowing into the capacitor C _c , i _p is the current in the primary side of the transformer, i _Lm is the current in the magnetizing inductance L _m , i T _is the primary side of the ideal transformer Current, i _s1 is the current flowing through the switch S ₁ , v _g1 is the control signal of the switch S ₁ , v _g2 is the driving signal of the switch S ₂ , i _s is the secondary current of the transformer, i _d1 is the current of the diode D _o1 , i _d2 is the current flowing into the diode D _o2 , i _Cr1 is the current flowing into the capacitor C _r1 , i _Cr2 is the current flowing into the capacitor C _r2 , i _ss is the current flowing into the output capacitor C _o and the load, i _o is the output current, U _o is the output voltage. In addition, the voltage parameters not marked in Fig. 6 are respectively: V _B is the voltage across the capacitor C _B , V _c is the voltage across the capacitor C _c , V _Cr1 is the voltage across the capacitor C _r1 , V _Cr2 is the voltage across the capacitor C _r2 Terminal voltage, V _o is the voltage across the output capacitor C _o .

In the primary side control circuit, EA is an error amplifier, Ce and _Re are compensation capacitors and compensation resistors , Vref is a reference voltage, _vsaw is a sawtooth wave signal _{, ve} is _an error signal, and COMP is a comparator.

In the secondary side constant current/constant voltage control circuit, EA1 and EA2 are error amplifiers, D _c1 and D _c2 are diodes, C _e1 and R _e1 are the compensation capacitors and compensation resistors of EA1, respectively, C _e2 and R _e2 are the compensation capacitors of EA2, respectively. Compensation capacitor and compensation resistor, U _ref1 and U _ref2 are the reference voltages of EA1 and EA2 respectively, u _e1 and u _e2 are the error signals of EA1 and EA2 respectively, u _rs1 and u _rs2 are the input signals of the reverse terminals of EA1 and EA2 respectively, k ₁ U _o is k ₁ times the output voltage, and k ₂ I _o is k ₂ times the output current. The secondary side control realizes constant voltage/constant current control, specifically: one is the output voltage error amplifier EA1, and the other is the output current error amplifier EA2. It can be known from the working principle of the control loop that when the output voltage is lower than the controlled target voltage of the constant voltage output loop, the error output u _e1 of EA1 is high level, at this time D _c1 is reversely cut off, and the constant voltage output control loop is Failure, the constant current output error amplifier EA2 works normally, and the output is constant current control; on the contrary, when the output current is lower than the target control current of EA2, the error output of EA2 is high level, at this time, D _c2 is reversely cut off, and the constant current output The control loop fails, the constant voltage output error amplifier EA1 works normally, and the output is constant voltage control. Through the above control, constant current/constant voltage control can be realized.

In addition, the control loop also includes: optocoupler isolation module, dead zone and drive circuit.

Further, the converter can be expanded into multiple outputs. For example, as shown in Figure 7, it is also possible to expand a single output to three current sharing outputs by adding a diode and two output capacitors. Specifically, the primary circuit of the transformer T remains unchanged, and the secondary side of the transformer T One end is connected to the anode of D _o1 and the resonant capacitor C _r1 , the other end of the resonant capacitor C _r2 ; the other end of the resonant capacitor C _r1 is connected to the anode of D _o3 and the cathode of D _o2 ; the positive pole of the output capacitor C _o1 is connected to the cathode of D _o3 , and the negative pole is connected to the cathode of D o3. Connect the other end of the secondary side of the transformer T, the output load R _o1 is connected to both ends of the output capacitor C _o1 ; the positive pole of the output capacitor C _o2 is connected to the resonant capacitor C _r2 , the negative pole is connected to the anode of D _o2 , and the output load R _o2 is connected to the output capacitor Both ends of C _o2 ; the negative pole of the output capacitor C _o3 is connected to the resonant capacitor C _r2 , the positive pole is connected to the cathode of D _o1 , and the output load R _o3 is connected to both ends of the output capacitor C _o3 . In the figure, 1:n is the turns ratio of the primary side of the transformer to the number of turns on the secondary side, L _m is the excitation inductance of the transformer, L _r is the leakage inductance of the transformer, C _c and S are the active clamp capacitor and the active clamp, respectively. bit switch. In Figure 7, T is the transformer, C _o is the output capacitance of the single output, and Ro is the _load of the single output.

The following takes a single output as an example to analyze its working process in detail. The topology and control circuit of the single-channel LED drive circuit are shown in Figure 6.

As shown _in Figure ₈ , the BuckPFC cell operates only when the input voltage |vin| is higher than the voltage _VB of the capacitor CB . In the figure | i _in | is the input current after full-wave rectification, and β is the duration of interval I. When the proposed converter works in steady state, there are two different operating states: 1) θ < ωt < π - θ (interval I); and 2) 0 < ωt < θ and π - θ < ωt < π ( interval II), as shown in Figure 8, where θ can be calculated by the following formula:

(1)

(1)

Among them, V _m is the peak value of the input voltage, and V _B is the voltage across the capacitor C _B.

1) θ < ωt < π -θ (interval Ⅰ)

Figure 9 shows the operating waveforms of the proposed PFC converter when it operates in DCM and θ < ωt < π -θ . In FIG. 9, t _on1 is the time when the switch S1 is turned on, t _{on2 is the} time when the switch S2 is _{turned on} , vg1 is the control signal of the switch S1 , _vg2 is _the driving signal of the switch S2 , vCr1 is the capacitor _{C r1} Voltage at both ends, v _Cr2 is the voltage across the capacitor C _r2 , i _LB is the current of the inductor LB , _{id is the current flowing into the DC-DC unit, i Lm is} the _current of the excitation inductor L _m , and i p _is the primary side of the transformer , i _s1 is the current flowing through the switch S ₁ , i _s is the secondary current of the transformer, T _r1 is the resonant period, i _ss is the current flowing into the output capacitor C _o and the load, i _B is the current flowing into the capacitor C _B , the horizontal axis t is the time axis, and Mode represents the mode. There are a total of 12 operating modes in one switching period T _s , and the corresponding equivalent circuits are shown in Figures 10a-10k. All parameters in Figures 10a-10k are consistent with those in Figure 6. Suppose, at rated power, T _dis > T _r2 /2, where T _dis is the discharge time of LB , and T _r2 is when the resonant circuit is composed of leakage inductance L r _, capacitance _{C c} , resonant capacitance C _r1 , C _r2 the resonance period. Furthermore, it is assumed that the magnetizing inductor current i _Lm < 0 and the secondary side current i _s = 0 before the start of Mode 1.

Mode 1 [ t ₀ ~ t ₁ ]: As shown in Figure 10a, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB , and i _B is the inflow The current of the capacitor C _B , i _d is the current flowing into the DC-DC unit, i _p is the current of the primary side of the transformer, i _Lm is the current of the excitation inductance L _m , i _T is the current of the ideal transformer primary side, i _s1 is the current flowing through The current of the switch S1 , i _s is the secondary current of the transformer, _{i d1} is the current of the diode D _o1 , i _d2 is the current of the diode D _o2 , i _Cr1 is the current flowing into the capacitor C _r1 , i Cr2 _is the current flowing into the capacitor C _r2 current, i _ss is the current flowing into the output capacitor C _o and the load, i _o is the output current, and the LEDs are the output load. _At time t0 , the switch S1 is turned _on , and the switch S2 is _turned off. Diodes D _B2 and D _{o1 conduct} and provide flow paths for i _LB and is , respectively. Diodes D _B1 , D _B3 and D _{o2 are} reverse biased. _The voltage across the buck inductor LB is | v _in | - VB , and the inductor current i _LB increases _linearly from zero:

(2)

(2)

Where t is the time variable, | v _in | is the input voltage after full-wave rectification, V _B is the voltage across the capacitor C _B , and L _B is the inductance of the Buck inductor.

At the same time, the magnetizing inductance L _m begins to charge C _B , the voltage across L _m is V _B , and i _Lm rises linearly:

(3)

(3)

where i _Lm ( t ₀ ) is the value of i _Lm at time t ₀ .

The resonant current i _s starts to flow through the secondary side, and the resonant circuit consists of the leakage inductance L _r and the resonant capacitors C _r1 and C _r2 , and i _s can be expressed as:

(4)

(4)

In the formula: v _Cr1 ( t ₀ ) is the voltage at both ends of C _r1 at time t ₀ , and n is the transformation ratio of the transformer T. Among them, the resonance angular frequency ω _r1 and impedance Z _r1 of the resonant circuit are respectively:

(5)

(5)

According to Kirchhoff's law, the transformer primary current i _p , the current i _B flowing through the capacitor C _B and the current i _s1 flowing through the active switch S ₁ can be expressed as:

(6)

(6)

where id is the current flowing into the DC-DC unit from the _BuckPFC unit; i _Lm is the current flowing through the transformer magnetizing inductance L _m ; n is the transformation ratio of the transformer T; i _s is the secondary current of the transformer; i _LB is the current flowing through the transformer T _Inductor LB current.

The currents i _Cr1 and i _Cr2 flowing through the two resonant capacitors have the following relationship:

(7)

(7)

where v _Cr1 and v _Cr2 are the voltages across the resonant capacitors C _r1 and C _r2 respectively, V _o is the voltage across the output capacitor C _o , and d is the differential sign. From (7), i _Cr1 = i _Cr2 can be obtained. Secondary side current i _s = i _Cr1 + i _Cr2 . Therefore, the current i _ss flowing through the output capacitor can be expressed as:

(8)

(8)

where i _Cr2 is the current flowing through the resonant capacitor C _r2 ; i _s is the secondary current of the transformer. Mode 1 ends when i _p increases from a negative value to zero.

Mode 2 [ t ₁ ~ t ₂ ]: As shown in Figure 10b, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB , and id is the _inflow The current of the DC-DC unit, i _p is the current of the primary side of the transformer, i _s is the current of the secondary side of the transformer, and the LEDs are the output load. At time t1 , the primary side current _{ip increases} to zero. The secondary side still resonates as it did in mode 1. Therefore , id , _{ip and} is _continue to increase. Mode 2 ends at time t _{2 when} id rises to be equal to i _LB.

Mode 3 [ t ₂ ~ t ₃ ]: As shown in Figure 10c, the current parameters in the figure are: | v _in | is the input voltage after full-wave rectification, i _B is the current flowing into the capacitor C _B , and i _d is The current flowing into the DC-DC unit, i _p is the current of the primary side of the transformer, i _Lm is the current of the excitation inductor L _m , i _s1 is the current flowing through the switch S ₁ , and the LEDs are the output loads. At time t ₂ , the capacitor current i _B is equal to zero. As i _d increases, i _B will change its flow direction. Therefore, DB3 _conducts and provides _a flow path for iB , while DB2 is _reverse biased. CB starts to supply energy to the secondary _side . The secondary side still resonates as analyzed in Mode 1. In mode 3, the current i _s1 flowing through S ₁ is equal to i _p . i _Lm continues to increase linearly. Mode 3 ends at time t ₃ when i _Lm increases from a negative value to zero.

Mode 4 [ t ₃ ~ t ₄ ]: As shown in Figure 10d, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB , and id is the _inflow The current of the DC-DC unit, i _Lm is the current of the excitation inductor L _m , and the LEDs are the output load. _At time t3 , the excitation inductor current i _Lm continues to increase. Other working states are the same as mode 3. _{Mode 4 ends when id changes from greater than iLB to} equal to _iLB .

modal 5[t ₄~t ₅]: As shown in Figure 10e, the current parameters in the figure are: |v _in| is the input voltage after full-wave rectification,i _Bfor the inflow capacitorC _Bthe current,i _LBis inductanceL _Bthe current,i _dis the current flowing into the DC-DC unit,i _s1to flow through the switchS ₁the current,i _sis the secondary current of the transformer,LEDsfor the output load. existt ₄time, capacitor currenti _B=i _d-i _LBequal to zero. along withi _LBincrease,i _Bwill change its flow direction.D _B2turned on and isi _Bprovide a flow path,D _B3reverse bias. In this modal, i _s1equali _LB. Other working states are the same as Mode 4. wheni _sresonates to zero,D _o1Zero current shutdown, mode 5 ends.

Mode 6 [ t ₅ ~ t ₆ ]: As shown in Figure 10f, the current parameters in the figure are: | v _in | is the input voltage after full-wave rectification, i _Lm is the current of the excitation inductor L _m , and the LEDs are the output load. _At time t5 , D _o1 is turned off, and S ₁ is still turned on. i _Lm continues to increase linearly. The output capacitor C _o discharges to provide power to the LED load. When the switch tube S1 is turned off, the switch tube S2 is turned _on , and the mode ₆ ends.

Mode 7 [ t ₆ ~ t ₇ ]: As shown in Figure 10g, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB , and i _B is the inflow The current of the capacitor C _B , i _d is the current flowing into the DC-DC unit, _ic is the current flowing into the capacitor C _c , i _p is the current of the primary side of the transformer, i _Lm is the current of the magnetizing inductance L _m , i s _is the transformer Secondary current, i _Cr1 is the current flowing into the capacitor C _r1 , i _ss is the current flowing into the output capacitor C _o and the load, and the LEDs are the output load. _At time t6 , the switch S2 is turned _on , and the switch S1 is _turned off. Diodes D _B1 and D _{o2 conduct} and provide flow paths for i _LB and is , respectively, and diodes D _B2 , D _B3 and D _o1 are all reverse biased. _The voltage across LB is VB and i _{LB decreases} linearly:

(9)

(9)

Among them, i _LB ( t ₆ ) is the value of i _LB at time t ₆ .

At the same time, part of the energy stored in L _m is transferred to C _c , and the other part is transferred to the secondary side. The voltage across Lm is Vc , _which is the voltage _across capacitor _{Cc .} i _Lm decreases linearly:

(10)

(10)

where i _Lm ( t ₆ ) is the value of i _Lm at time t ₆ .

The resonant current i _s starts to flow through the secondary side. At this time, different from mode 1, the resonant circuit is composed of leakage inductance L _r , capacitor C _c and resonant capacitors C _r1 and C _r2 . _Is can be expressed as:

(11)

(11)

The resonant angular frequency ω _r2 and impedance Z _r2 of the resonant circuit are respectively

(12)

(12)

Among them, L _r is the leakage inductance, n is the turns ratio, and C _r1 , C _r2 , and C _c are the inductances of the capacitors, respectively.

According to Kirchhoff's law, the current i _p on the primary side of the transformer can be expressed as:

(13)

(13)

Where ic is the current flowing through the capacitor C _c , i _Lm is the current flowing through the transformer magnetizing inductance, and i _s is the current on the secondary side of the transformer _.

The current i _ss flowing through C _o can be expressed as:

(14)

(14)

Mode 7 ends when i _p falls from a positive value to zero.

Mode 8 [ t ₇ ~ t ₈ ]: As shown in Figure 10h, the current parameters in the figure are: | v _in | is the input voltage after full-wave rectification, i _p is the current on the primary side of the transformer, and i _s is the transformer Secondary current, LEDs are output loads. At time t7 , i _{p is} reduced to zero. The secondary side still resonates in the mode of operation in mode 7. As ip and is _continue to decrease , _{ip will} change its direction. Mode 8 ends when i _p goes from negative to zero.

Mode 9 [ t ₈ ~ t ₉ ]: As shown in Figure 10g, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB , and i _B is the inflow The current of the capacitor C _B , i _d is the current flowing into the DC-DC unit, _ic is the current flowing into the capacitor C _c , i _p is the current of the primary side of the transformer, i _Lm is the current of the magnetizing inductance L _m , i s _is the transformer Secondary current, i _Cr1 is the current flowing into the capacitor C _r1 , i _ss is the current flowing into the output capacitor C _o and the load, and the LEDs are the output load. _At time _t8 , i p increases to zero. The equivalent circuit and theoretical analysis of this mode are identical to those of Mode 7. When i _s resonates to zero, D _o2 is turned off at zero current, and mode 9 ends.

Mode 10 [ t ₉ ~ t ₁₀ ]: As shown in Figure 10i, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB _, and i Lm is the excitation The current of the inductor L _m , the LEDs are the output load. _At time t9 , D _o2 is turned off. D _o1 and S ₂ are still open. iLB and _iLm continue to decrease _linearly . The output capacitor C _o discharges to supply power to the LED load. When i _LB decreases to zero, D _o1 is turned off at zero current, and mode 10 ends.

Mode 11 [ t ₁₀ ~ t ₁₁ ]: As shown in Figure 10j, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB _, and i Lm is the excitation The current of the inductor L _m , the LEDs are the output load. _At time t10 , DB1 is _turned off. i _Lm continues to decrease linearly. Mode 11 ends when i _Lm decreases to zero.

Mode 12 [ t ₁₁ ~ t ₁₂ ]: As shown in Figure 10k, the current parameters in the figure are: | v _in | is the input voltage after full _- wave rectification, i _LB is the current of the inductor LB _, and i Lm is the excitation The current of the inductor L _m , the LEDs are the output load. _At time t11 , i _Lm decreases to zero. i _Lm continues to decrease in the manner analyzed in Mode 7. When the switch S2 is turned off, the switch S1 is turned _on again, and _one switching cycle ends.

As shown in Figure 11, in the interval θ < ωt < π - θ , since i _LB varies with the input voltage | v _in |, when i _LB varies, mode 4, mode 5 and mode 6 There will be different situations. The above analysis of mode 4, mode 5 and mode 6 is carried out by taking interval C as an example. In fact, the analysis of intervals A and B is similar to that of interval C.

2) 0 < ωt < θ and π -θ < ωt < π (interval II)

As shown in Figure 14, when | v _in | is lower than VB , _{iLB is} always zero. Figure 12 shows the main waveforms of the proposed PFC converter when 0 < ωt < θ and π - θ < ωt < π. In Fig. 12, t _on1 is the turn _- on time of switch S1 , _{t on2} is the turn _- on time of switch S2 , _vg1 is the control signal _of switch S1 , _vg2 is the drive signal of switch _S2 , iLB is the inductance _LB , i _d is the current flowing into the DC-DC unit, i _Lm is the current of the excitation inductance L _m , i _p is the current of the primary side of the transformer, i _s1 is the current flowing through the switch S ₁ , and i _s is the secondary side of the transformer Current, T _r1 is the resonance period, _iss is the current flowing into the output capacitor C _o and the load , i B _is the current flowing into the capacitor C _B , the horizontal axis t is the time axis, and Mode represents the mode. In one switching cycle, there are 9 modes, and their corresponding equivalent circuit diagrams are shown in Fig. 13a-Fig. 13d. All parameters in Figures 13a-13d are consistent with Figure 6 .

Mode 1 [ t ₀ ~ t ₁ ]: As shown in Figure 13a, the current parameters in the figure are: | v _in | is the input voltage after full-wave rectification, i _B is the current flowing into the capacitor C _B , and i _d is The current flowing into the DC-DC unit, i _p is the current of the primary side of the transformer, i _s1 is the current flowing through the switch S ₁ , i _Lm is the current of the excitation inductance L _m , i _s is the secondary current of the transformer, and the LEDs are the output load . At time t ₀ , the switch S1 is turned _on , and the switch S2 is _turned off. The working mode of the _magnetizing inductor current i _Lm and the secondary side current is is the same as Mode 1 in 3.3.1. The current i _s1 flowing through S ₁ is zero in this mode. i _p , id and i _{B can} be expressed as:

(15)

(15)

Among them, _{id is the current flowing into the DC-DC unit, i B is the current flowing through the capacitor CB, i Lm} is the _{current flowing} through _the transformer magnetizing inductance , i s _is the secondary side current of the transformer, and n is the turns ratio.

The other works are the same as mode 1 in θ < ωt < π -θ (interval I). Mode 1 ends when i _p increases from a negative value to zero.

Mode 2 [ t ₁ ~ t ₂ ]: As shown in Figure 13b, the current parameters in the figure are: | v _in | is the input voltage after full-wave rectification, i _B is the current flowing into the capacitor C _B , and i _p is The current of the primary side of the transformer, i _s1 is the current flowing through the switch S ₁ , i _Lm is the current of the excitation inductor L _m , and the LEDs are the output load. As i _p increases, i _B will be greater than zero. CB starts to supply power to the secondary _side . The secondary side continues to resonate. i _s1 is equal to i _p . i _Lm continues to increase linearly. Mode 2 ends when i _Lm increases from a negative value to zero.

Mode 3 [ t ₂ ~ t ₃ ]: As shown in Figure 13c, the current parameters in the figure are: | v _in | is the input voltage after full-wave rectification, i _Lm is the current of the excitation inductor L _m , and i _s is Transformer secondary current, LEDs are output loads. As i _Lm increases linearly, its flow direction will change. Others work the same as modal 2. When i _s increases to zero, D _o1 is turned off at zero current, and mode 3 ends.

Mode 4 [ t ₃ ~ t ₄ ]: As shown in Figure 13d, the current parameters in the figure are: | v _in | is the input voltage after full-wave rectification, _id is the current flowing into the DC-DC unit , and i _p is the current of the primary side of the transformer, i _s1 is the current flowing through the switch S ₁ , i _Lm is the current of the excitation inductor L _m , and the LEDs are the output loads. i _Lm continues to increase linearly. C _B charges L _m . i _{s1 ,} id and i _p are all equal to i _Lm . C _o provides power to the LED load. When S1 is turned off and S2 is turned _on , Mode ₄ ends.

Respectively remove the loops composed of L _B , C _B and D _B1 in mode 7, mode 8 and mode 9 in 1) θ < ωt < π-θ (interval Ⅰ), respectively, and we can get the results in 0 < Equivalent circuit and circuit analysis of Mode 5, Mode 6, and Mode 7 for ωt < θ and π -θ<ωt < π.

Mode 8 and Mode 9 are the same as Mode 11 and Mode 12 in 1) θ < ωt < π -θ (interval I), respectively.

Figure 14 shows the input voltage, current waveform and Buck inductor current waveform in one power frequency cycle and switching cycle. In the figure, β = π - 2 θ is the interval length, DT _s is the conduction time of the switch S ₁ in FIG. 6 , dT _s is the conduction time of the diode D _B1 in FIG. 6 , and t is the time. The average value of the input current i _in-avg in one switching cycle can be expressed as:

(16)

(16)

Among them, D is the duty cycle, T _s is the switching period, L _B is the inductance of the Buck inductor, V _B is the voltage across the capacitor C _B , and | v _in | is the input voltage after full-wave rectification.

From equation (16), it can be seen that high power factor can be achieved when D and T _s remain constant for half the power frequency cycle and VB is _low enough.

When the proposed PFC converter operates in DCM, the maximum stress _of active switch S1 is:

(17)

(17)

Among them, V _m is the peak value of the input voltage, V _c is the voltage across the capacitor C _c , and V _o is the output voltage.

The maximum stress of switch tube S ₂ is:

(18)

(18)

At the same time, the LED driver proposed by the present invention eliminates the influence of the voltage ripple of the intermediate energy storage capacitor on the output voltage through the voltage fast loop control, so the LED driver can realize low ripple current output.

Table 1 shows the experimental parameters of the low output ripple power factor correction converter.

Table 1 Experimental parameters of low output ripple power factor correction converter

Figure 15 shows the power factor (PF) and efficiency of the prototype. In the figure, the ordinate PF is the power factor, and the abscissa v _in (rms) [V] is the rms value of the input voltage, in volts. It can be seen that the power factor of the prototype is higher than 0.956 in a wide input voltage range. It can also be seen that the efficiency of the prototype can reach up to 91.08%. Therefore, the converter can achieve high power factor and high efficiency.

Figure 16 shows the harmonic measurements of the input current at 220Vac input voltage. It can be seen that each current harmonic (up to the 15th order) is smaller than that required by IEC61000-3-2 Class C, and there is a certain margin.

According to the above analysis, the converter proposed by the present invention can achieve high power factor, high efficiency, low output ripple, small active switch voltage stress, and is easy to expand into multiple outputs.

The invention not only realizes power factor correction and low output ripple, but also solves the problems of large voltage stress and voltage spikes in the main switching tube of the traditional integrated converter, and can recover the leakage inductance energy of the transformer, thereby further improving the overall performance of the converter. efficiency. The output of the present invention is easy to expand into multi-channel current sharing output, which is very suitable for LED driver. The invention has the advantages of high efficiency, high power factor, low output ripple, and easy expansion into multiple outputs. In addition, the present invention is an isolation structure, which is more secure.

Claims (4)

1. a low output ripple power factor correction converter, is characterized in that, comprises the DC-DC converter of rectifier bridge, filter, Buck PFC converter and rear stage, and control circuit thereof; Be specially: input filter inductance L _f and the input filter capacitor C _f are connected in series with the output end of the rectifier bridge D _b ; one side of the input filter capacitor C _f is connected between the input filter inductor L _f and the Buck inductor L _B , and the other side is connected to the diode rectifier bridge The lower output terminal of D _b ; diodes D _B1 and D _B3 are connected in series and parallel to both ends of the input filter capacitor C _f , the anode of D _B3 is connected to the lower output terminal of D _b , and the cathode of D _B1 is connected between L _f and Buck inductor L _B ; _The positive stage of the intermediate energy storage capacitor _CB is connected to one end of the Buck inductor LB , and the negative electrode is connected between D _B1 and D _{B3 ;} the anode of the diode D _B2 is connected to the negative electrode of the intermediate energy storage capacitor _CB , and the cathode is connected to the active switch S 1 One end of the primary side of the transformer T is connected to the positive stage of the intermediate energy storage capacitor C _B , and the other end is connected to the drain _of the active switch S1 and the source of the active switch S2 _; the active _switch S2 and the energy storage The capacitor C _c is connected in series and parallel to both ends of the transformer T. The drain of the active switch S ₂ is connected to the energy storage capacitor C _c , and the gate is connected to the control loop; the source of the active switch S ₁ is connected to the diode rectifier bridge D _{b The} lower output end, the drain is connected to the source of S2 , and the gate is connected to the control loop; one end of the secondary side of the transformer T is connected to the anode of D _o1 and the cathode of D _o2 , and the other end is connected to two series resonant capacitors C Between _r1 and C _r2 ; the cathode of _Do1 is connected to the resonant capacitor C _r1 and the output capacitor C _o ; the resonant capacitor C _r1 and C _r2 are connected in series and connected to the cathode of D _o1 and the anode of D _o2 ; the output capacitor C _o is connected in parallel It is connected to both ends of the resonance capacitor C _r1 and C _r2 after being connected in series; the output load is connected to both ends of the output capacitor C _o . 2. a kind of low-output ripple power factor correction converter according to claim 1, is characterized in that, by adding a diode and two output capacitors, single-channel output is expanded into three-way current-sharing output, specifically: The primary circuit of the transformer T remains unchanged. One end of the secondary side of the transformer T is connected to the anode of D _o1 and the resonant capacitor C _r1 , and the other end of the resonant capacitor C _r2 ; the other end of the resonant capacitor C _r1 is connected to the anode of D _o3 and D _o2 Cathode; the positive pole of the output capacitor C _o1 is connected to the cathode of D _o3 , the negative pole is connected to the other end of the secondary side of the transformer T, the output load R _o1 is connected to both ends of the output capacitor C _o1 ; the positive pole of the output capacitor C _o2 is connected to the resonant capacitor C _r2 , The negative pole is connected to the anode of D _o2 , the output load R _o2 is connected to both ends of the output capacitor C _o2 ; the negative pole of the output capacitor C _o3 is connected to the resonant capacitor C _r2 , the positive pole is connected to the cathode of D _o1 , and the output load R _o3 is connected to the output capacitor C _o3 . end. _3. a kind of low output ripple power factor correction converter according to claim ₁ , it is characterized in that, active switch S1 and S2 are complementary conduction, specifically _: S2 _first before S1 conduction Turn off, both active switches S ₁ and S ₂ are turned off at this time; after a short dead time, S ₁ is turned on, and S ₂ remains off; when S ₁ receives the turn-off signal, it turns off immediately, and S ₂ continues to hold Turn off; after a short dead time, S ₂ is turned on, and S ₁ is kept off; the dead time is generated by the driver chip IR2111, specifically: the control signal generated by the control circuit is input to the driver chip IR2111, and then the IR2111 generates the band independently. The active switches S ₁ and S ₂ are driven respectively by two complementary and symmetrical driving signals with dead zones. 4. a kind of low output ripple power factor correction converter according to claim 1 is characterized in that, there are two error amplification control loops in the secondary side control circuit, one is the output voltage error amplifier EA1, and the other is the output Current error amplifier EA2; according to the working principle of the control loop, when the output voltage is lower than the controlled target voltage of the constant voltage output loop, the error output u _e1 of EA1 is high level, at this time D _c1 is reversely cut off, and the constant voltage The voltage output control loop fails, the constant current output error amplifier EA2 works normally, and the output is constant current control; on the contrary, when the output current is lower than the target control current of EA2, the error output of EA2 is high, and D _c2 is reversed at this time. When it is cut off, the constant current output control loop fails, the constant voltage output error amplifier EA1 works normally, and the output is constant voltage control; through the above control, constant current/constant voltage control can be realized.

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